Aircraft navigation computer

ABSTRACT

An all-electronic solid-state airborne analog computer, utilizing the navigation signals available from a VOR receiver and a DME receiver to develop linear coordinate signals continuously indicating the aircraft position and direction of movement relative to a ground station or to an arbitrarily selected waypoint within the operating range of the ground station.

United States Patent 7 Anthony [4 Apr. 25, 1972 s41 AIRCRAFT NAVIGATIONCOMPUTER 3,495,241 2/1970 Perkins ..343/o x 3,534,399 10/1970 Hirsch..343/l07 X [72] Inventor: Myron L. Anthony, La Grange, [IL [73]Assignees: Thomas E. Dorn, Clarendon Hill; Statisti- Primary Examiner-T.H. Tubbesing cal Services, inc., Chicago, Ill. part in-Atmrney-KinzenDorn and Zickert terest to each 221 Filed: Aug. 18, 1969i571 ABSTRACT [2]] Appl. No.: 851,028 An all-electronic solid-stateairborne analog computer, utilizing the navigation signals availablefrom a VOR receiver and a DME receiver to develop linear coordinatesignals continu- [52] U.S.CI. ..343/6 R,343/ 106 R ousl indicatin theaircraft osmon and direction of move [51] ...G0ls 9/56 G015 l/44 y g P[58] Field 0 Search 34376 R 106 R ment relative to a ground station orto an arbitrarily selected waypoint within the operating range of theground station.

[56] References Cited 14 Claims, 11 Drawing Figures UNITED STATESPATENTS 3,581,073 5/1971 Visher ..343/6 R |8 voR Recrswmz ll HI-PASS l4mscmmrm 10K Fl 135R I f FILTER I RF xr'mo Dfi ra'roa lb as l? STAGES36h:

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T Pcos e R$m4 Psm. s 72 Y1, P cos 9 Y F 6.3 R+c0s VORTAC STATIQN X=R sin(P l X! Psine Y-R cos (I) AIRCRAFT POSlTlON Inventor ,Mgrozz Lfinfhog I32 x gfl pmadzickmt fl'fiornegs PATFNTEDAPRZ me I I 3,659,291

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SHEET 6 BF 6 FROM DME 3 4, RECWER l2 7 F 8 Fl 6.9 40v WAY POINT SELECTOR72 FIGJO 4 R os+Pcose 4&2 M Q 413 5UMMIHG Inventor AMPLIFIER Ma rm 66 Pe5 K wfimw Zu/ke/fl:

AIRCRAFT NAVIGATION COMPUTER CROSS REFERENCES TO RELATED APPLICATIONSThe computer of the present invention-preferably is used in conjunctionwith a VOR receiver of the kind described and claimed in the co-pendin'gapplication of Myron L. Anthony, Ser. No. 7 1 3,786, filed Mar. 18,1968, now abandoned and superseded by application Ser. No. 54,778, filedJuly 14, I970. The preferred forms of the computer employ constant slewrate circuits described and claimed in the co-pending application ofMyron L. Anthony, Ser. No. 11,399, filed Feb. 16, 1970.

BACKGROUND OF THE INVENTION In conventional aircraft navigationinstrumentation, the bearing of the aircraft with respect to a givenground station transmitter (VORTAC) is determined by bearing signalsreceived and interpreted by a VOR receiver, whereas the distanceinformation pertinent to the position of the aircraft relative to thesame ground station is derived from the distance signals developed by aDME receiver. The bearing and distance of the aircraft with respect tothe ground station may be displayed on separate instruments. On theother hand, these two different types of information may be combined ina single situation display.

In recent years, increasing attention has been given to course linecomputers for combining the bearing and distance information in anintegrated display to enable the pilot to determine more rapidly andconveniently the present location and movement of the aircraft withrespect to the ground navigation station. A course line computer can beof substantial value in increasing the utility of existing groundfacilities and in presenting the navigational position information tothe pilot in a more effective manner than is possible with separatedisplays of bearing and distance information. Moreover, the course linecomputer makes it possible, in many instances, to select an arbitraryway point and to display bearing and distance information with respectto the waypoint instead of relative to the ground station. Thiscapability makes it practical and convenient to fly multiple parallelpaths that do not-intersect'the location of the ground station,increasing the air traffic that can be safely and effectively guidedfrom an existing ground station by an order of magnitude.

But known course line computers have presented a number of difficultiesin their construction and operation. In one known course line computer,as described in Perkins U.S. Pat. Nos. 2,801,051 and 3,034,724,resolution of the bearing and distance data into linear coordinatesignals suitable for use in an effective integrated position display wasachieved with the use of precision rotary DC potentiometers. Thatconstruction afforded an effective and accurate display only if thepotentiometers were wound with extreme accuracy, an accuracy that proveddifficult to achieve, and even more difficult to duplicate.

In more recent proposals for course line computers, particularly asdescribed in Perkins and Anthony US. Pat. No. 3,4l4,90l, AC resolvershave been utilized, instead of the earlier DC potentiometers, as a meansfor combining the distance and bearing data signals to develop usablelinear coordinate signals. Resolver systems impose considerably lessdifficulty, in regard to precision manufacture and matching ofcomponents, when compared with DC potentiometers, but usually requirerelatively complex and expensive servo systems. Moreover, the resolversmust still be manufactured to relatively high precision standards andare also rather expensive. Furthermore, course .line computers.utilizing AC resolvers continue to present substantial technicalproblems with respect to the required intermodulation of distance'andbearing data particularly with respect to distortion of the bearinginformation. a I

SUMMARY OF THE INVENTION It is a principal object of the invention toprovide a new and improved analog position computer for an aircraftnavigation system that is an all-electronic solid state device and thatrequires no rotary potentiometers or resolvers in its operation.

Another object of the invention is to provide a new and improvedall-electronic solid state analog aircraft navigation position computerthat effectively and inherently eliminates or minimizes the possibilityof introduction of error in the computer operation.

Another object of the invention is to provide a new and improvedall-electronic analog position computer for an aircraft navigationsystem that is substantially simpler and less expensive than digitalcomputer apparatus but entails no efiective loss in speed or accuracy.

A further object of the invention is to provide a new and improved solidstate analog aircraft position computer that effectively modifies thebearing and distance data from a conventional ground station to enablethe pilot to fly the aircraft on a given course relative to a waypointthat is entirely separate from the ground station.

Another object of the invention is to provide a new and improved solidstate analog aircraft position computer that is capable of using raw VORreceiver data without the introduction of errors often introduced byconventional VOR detection systems.

A specific object of the invention is to provide a new and improvedsolid state aircraft navigation computer that is simple and inexpensivein construction, and requires a minimum of cockpit space.

Accordingly, the invention is directed to an analog solidstate positioncomputer for use in an aircraft navigation system of the kind includinga receiver for developing first and second data signals of given fixedfrequency varying in phase angle 4: relative to each other, inaccordance with one position parameter, and a receiver for developing athird data signal representative of another position parameter, relativeto a ground station transmitter. The computer comprises means forutilizing the first data signal to generate a first intermediate signal,at the given frequency, of substantially rectangular wave form.Modulating means are provided for amplitude modulating the second datasignal in accordance with the third data signal to generate a secondintermediate signal of amplitude R, at the given frequency, having asinusoidal wave form. A first product detector multiplies the twointermediate signals together to generate a first ground stationcoordinate signal R cos A second product detector multiplies theintermediate signals together, with one intermediate signal shifted 90in phase, to generate a'second ground station coordinate signal R sin11:.

DESCRIPTION OF THE DRAWINGS usable in the embodiment of FIG. 4;

FIG. 8A illustrates the output signals from the circuit of FIG. 8; and

used in the system of FIG. 4..

FIGS. 9 and I0 illustrate additional circuits that may be DESCRIPTION OFTHE PREFERRED EMBODIMENTS FIG. 1 illustrates an all-electronic solidstate analog aircraft navigation position computer 10 constructed inaccordance with one embodiment of the present invention and utilizingthe signals developed by a VCR receiver 11 and a DME receiver 12. Itshould be understood that the block diagram of FIG. 1 has beensubstantially simplified, in comparison to the overall circuitry of thecomputer and'the two receivers, with only the fundamental operatingcircuits illustrated.

The VOR receiver 11, as shown in FIG. 1, may be of substantiallyconventional construction, preferably with the conventional output stageomitted. Receiver 11 includes an antenna 13 coupled to the initialstages 14 of the receiver, including the usual radio frequency,intermediate frequency, and detector stages. The output of circuit 14 iscoupled to two signal channels 15 and 16 in receiver 1 1, channellSbeing the reference signal channel and channel 16 being the variablesignal channel.

The reference signal channel 15 in receiver 11 may comprise a high-passfilter 17 having its output coupled to a discriminator 18. The output ofdiscriminator 18 is in turn coupled to a filter 19. The output of filter19, at terminal 21, is the constant-phase 30 hz. reference signal of theVCR receiver.

Signal channel 16 in receiver 11 comprises a low pass filter 22. Theoutput of filter 22 is connected to a filter 23. The output of filter23, at terminal 24, is the 30 hz. variable phase signal of the VCRreceiver.

As noted above, receiver 1 1 may be entirely conventional inconstruction. It is much preferred, however, that the VCR receiver beconstructed with integrating filters or other constant slew rate filtersfor the filter circuits 19 and 23, instead of the passive filters thathave conventionally been used in receivers of this kind. Integratingfilters make it possible to eliminate many of the errors that mightotherwise be introduced in the receiver operation, due to the inherentamplitude sensitivity and phase shift of conventional passive filters.Moreover, discriminator 18 is preferably a product discriminator of thekind described in Anthony U.S. Pat. No. 3,024,419. Furthermore, it ismost desirable to utilize electrical signal outputs from the VORreceiving circuits as the computer inputs, without resort to any outputservo or other electromechanical output device. Preferred constructionsfor the circuits of the VCR receiver 11 are described in substantialdetail in the aforementioned co-pending application of Myron L. Anthony,Ser. No. 713,786.

DME receiver 12 may be conventional in construction; the internalcircuits of the DME receiver have not been shown in the drawing. Theoutput of receiver 12, at terminal 25, is a DC signal that varies inamplitude in accordance with the distance of the aircraft from theground station transmitter to which receiver 12 is tuned.

In computer 10, the variable bearing data signal from output terminal 24of VOR receiver 11 is coupled to one input of a limiter-modulatorcircuit 26. Modulator 26 also has a second input that is connected tothe output 25 of the DME receiver 12. The output of modulator 26 iscoupled to a filter 27 comprising a first integrator 28 and a secondintegrator 29 connected in series with each other. The output of filter27 is coupled to one input of a first product detector comprising adifferential amplifier 31. The output of filter 27 is also connected toone input of a second product detector comprising a differentialamplifier 32.

The reference signal from output terminal 21 of VOR receiver 11 isconnected, in computer 10, to the input of a limiter or other squarewave signal generator circuit 33. The output of limiter 33 is connectedto a second input of the differential amplifier 31. The output oflimiter 33 is also connected to the input of a phase shift circuit 34.Phase shifter 34, which has a phase shift of 90 at 30 hz., is connectedto a second input for the second product detector comprisingdifferential amplifier 32. The outputs of the two differentialamplifiers 31 and 32 are both connected to a display device 35.

In operation, signals received at antenna 13 of VOR receiver 11 aresupplied to the preliminary receiver stages 14. In stages 14, thereceived signals are amplified and detected in the usual manner,producing an output signal that is supplied In channel 16, low-passfilter 22 effectively restricts the 7 signal supplied to filter 23 tothe 30 hz. variable bearing data signal. Extraneous noise is removedfrom the signal in filter 23 and the 30 hz. variable phase signal issupplied from output terminal 24 to limiter-modulator 26in computer 10.

The bearing data signal supplied to limiter-modulator 26 develops asquare wave output signal 42 having an amplitude that varies inaccordance with the input signal from receiver 12. Moreover, the phaseof signal 42 varies in accordance with the variations in the phase ofthe input signal 41 from receiver 1 1.

In filter 27, the square wave signal 42 is integrated twice, producingan output signal 43 of sinusoidal wave form. The dual integration doesnot change the phase or amplitude of signal 43, as compared with signal42. Thus, signal 43, which is supplied to the two product detectorscomprising differential amplifiers 31 and 32, varies in amplitude R inaccordance with the distance signal from receiver 12, and varies inphase in accordance with the variable phase signal from receiver 11.

The input signal to limiter 33, taken from output terminal 21 ofreceiver 11, is a sine wave signal 44. In limiter 33, which may comprisea conventional clipping amplifier, the reference 30 hz. signal 44 isconverted to a square wave signal 45; signal 45 is supplied to one inputof differential amplifier 31 and also to the input of phase shifter 34.The output of the phase shifter is a signal 46 of rectangular wave formhaving the same frequency as signal 45, but with a phase delay of 90.

Amplifier 31 operates as a product detector with respect to the twoinput signals 43 and 45 supplied thereto. The output of am lifier 31 isa DC signal which can be shown to have an amplitude and polarity thatvaries with changes in the amplitude R of the sinusoidal input signal 43and with changes in the cosine of the phase angle d of the square waveinput signal 45. Similarly, amplifier 32 functions as a product detectorwith respect to its two input signals 43 and 46. The output signal fromamplifier 32 is a DC signal having an amplitude R sin 4:. It will berecognized that the two output signals R cos d and R sin of the twoproducts detectors 31 and 32, respectively, are linear coordinatesignals that fully define the position of the aircraft with respect tothe ground station as regards both bearing and distance. These twosignals are supplied to the display device 35 to actuate the display andafford to the pilot a continuous indication of his position relative tothe ground station. Display device 35 may be substantially conventionalin construction; for example, a Sperry RDI00 radio direction indicator,with the usual auxiliary equipment,

may be employed.

From the foregoing description, it will be seen that computer 10 is anall-electronic solid state analog computer that requires no servos,precision potentiometers, resolvers, or other similar electro-mechanicalcomponents. The computer, when constructed with integrated circuits andother solid state devices, is extremely small in size and occupies aminimum of space. In fact, it can be readily fabricated as the outputstage of a VCR receiver with no need for enlargement of the receiverhousing. On the other hand, high accuracy is easily achieved in computer10, and there is little or no opportunity to introduce distortion intothe linear coordinate signals developed by the computer. The productdetectors 31 and 32 are of substantial value in preventing distortion inoperation of the computer. Moreover, the computer components arerelatively inexpensive, particularly in comparison with a digitalcomputer or with a conventional analog computer capable of performingcomparable operations and based upon servos, otentiometers, resolvers,or like electromechanical components.

FIG. 2 illustrates a somewhat more sophisticated analog computer 50constructed in accordance with another embodiment of the invention. Incomputer 50, the output terminal 24 of the VOR receiver apparatus 11 isconnected to the input of a zero crossing detector 51. The output ofcircuit 51 is connected to one input of an amplitude modulator 52.Modulator 52 has a second input connection from the output terminal 25of the DME receiver 12. Modulator 52 is a gated chopper circuit, theoutput from detector 51 being connected to the gate control and the DCsignal from receiver 12 constituting the signal that is chopped.

The output of modulator 52 is connected to the input of an integrator 28which is in turn connected to a second integrator 29 in a filter circuit27. The output of the integrator 29 is connected to a cosine detector31. The output of integrator 29 is also connected to a phase shiftercircuit 56 having its output connected to a sine detector 32. Each ofthe two detectors 31 and 32 is a differential amplifier functioning as aproduct detector as described above in connection with FIG. 1.

The reference output terminal 21 of receiver 11 is connected to theinput of a zero crossing detector 57. The output of detector 57 isconnected to a second input for product detector 31. The output ofcircuit 57 is also connected to a second input for product detector 32.

Computer 50 (FIG. 2) further comprises an oscillator 58 operating at arelatively low frequency, a frequency that is not harmonically relatedto the 30 hz. signals of VOR receiver 11 or to the power supplyfrequency of the aircraft. Assuming that the power supply of theaircraft operates at a nominal 400 hz., as is usually the case, thefrequency for oscillator 58 may be selected as 465 hz. as indicated inFIG. 2, although other frequencies can be used as desired. Oscillator 58has two output terminals 61 and 62. Terminal 61 provides an outputsignal of rectangular wave form and terminal 62 produces a correspondingfrequency and phase but of sinusoidal wave form.

The output terminal 61 of oscillator 58 is connected to one input of amodulator means 60 comprising an initial modulator circuit 63, which maycomprise a field-effect transistor chopper circuit. Modulator 63 has asecond input connection taken from the output of the cosine productdetector 31. The output of modulator 63 is supplied to an integrator 64which is in turn coupled to a second integrator 65, both included inmodulator means 60. The output of second integrator 65 is connected toone input of a summing amplifier 66.

' The square wave output terminal 61 of oscillator 58 is also connectedto one input of a second modulator means 70 comprising achopper-modulator 67 having a second input taken from the output of thesine product detector 32. The output of modulator 67 is connected to theinput of an integrator 68 which is in turn coupled to the input of asecond integrator 69, both included in modulator means 70. The output ofintegrator 69 is connected to one input of a summing amplifier 71.

Computer 50 further includes a way point selector 72. Way point selector72 has an input circuit connected to the sinusoidal signal outputterminal 62 of oscillator 68. The way point selector has two manualadjustment devices, a bearing adjustment 73 and a distance adjustment74. Way point selector 72, which is discussed in greater detailhereinafter, produces two output signals P sin 0 and P cos 0, determinedby the settings of the distance and bearing adjustments 73 and 74, thesesignals appearing at the output terminals 75 and '76 respectively. Theoutput signal P sin 0 from terminal 75'is supplied to a second input ofthe summing amplifier 71. The way point signal P cos 0 from terminal 76is supplied to the second input of summing amplifier 66. The outputs ofthe two summing amplifiers 66 and 71 are connected to a displayapparatus 35A illustrated as including adisplay instrument 35, a trackdeviation detector 91, a to-from detector 92, and a waypoint servomechanism 93.

It will be recognized that devices 91-93 correspond to conventionalauxiliary equipment for display device 35; the servo 93 corresponds tothe output servo apparatus of a DME receiver. The distance dataconnection from servo apparatus 93 to display device 35 includes aswitch 94 that can connect the display device to DME receiver 12 insteadof the way point servo. Detectors 91 and 92 each have an inputconnection from oscillator 58. In a preferred construction, a constantslew rate filter of the kind discussed hereinafter in connection withFIGS. 4 and 7, is interposed in the course deviation control circuitbetween detector 91 and display device 35.

The overall operation of computer 50 (FIG. 2) is somewhat substantialdifferences in the two computers; furthermore,

' computer 50 provides for the introduction of way point information notemployed in computer 10. The sinusoidalvariable phase 30 hz. signal 41derived from output terminal 24 of VOR receiver 11 is converted to acorresponding signal of rectangular wave form by the zero crossingdetector 51. This square wave signal 81 is modulated in amplitude inmodulator 52, producing an output signal 42A having the essentialcharacteristics of signal 42 as described above in connection withFIG. 1. As before, signal42A is integrated twice, in circuits 28 and 29,producing a sinusoidal intermediate signal 43A having a frequency andphase corresponding to that of the variable phase signal from the VORreceiver, but having an amplitude determined by the distance signal fromthe DME receiver. That is, the signal 43A appearing at the output ofintegrator 29 has an amplitude R corresponding to the amplitude of theDME signal and corresponds to the signal 43 developed in computer 10.

The constant phase reference signal 44 from VOR receiver terminal 21 isconverted to a square wave intermediate signal 45A by the zero crossingdetector 57. Signal 45A corresponds in all essential respects to signal45 of the previously described embodiment, the only difference being thetype of circuit employed to generate the square wave signal.

The two product detectors 31 and 32 function as before to develop linearcoordinate signals R cos d: and R sin (b respectively. The firstcoordinate signal R cos (b is supplied to modulator 63 and the secondcoordinate signal R sin d is supplied to modulator 67. In modulator 63,the first coordinate signal R cos d) is employed to modulate theamplitude of the 465 hz. square wave signal 82 from oscillator 58. Theoutput signal from modulator 63 is of rectangular wave form; it isintegrated in circuit 64 to produce a signal of triangular wave form andis further integrated in circuit 65 to produce a signal 83 of sinusoidalwave form that is one of the input signals to summing amplifier 66. Theother input signal to amplifier 66 is the sinusoidal way pointcoordinate signal 84 of amplitude P cos 0 from terminal 76 of way pointselector 72. In amplifier 66, the two input signals 83 and 84 arealgebraically added, producing a sinusoidal position signal 85, R cos d:P cos 6. This is a linear coordinate signal relating the presentposition of the aircraft to the way point selected by way point selector72, but based upon the navigation signals received by receivers 1 l and12, and is supplied to display device 35 in apparatus 35A.

The other coordinate signal for display apparatus 35A is developed inthe same manner. The DC output signal R sin d: from detector 32 isemployed, in modulator 67, to modulate the amplitude of the square wavesignal 82 from oscillator 58. The modulated signal is twice integratedto produce a sinusoidal ground station coordinate signal 86 that issupplied to one input of summing amplifier 71. In amplifier 71, signal86 is algebraically added'with the way pointsignal 87 from way pointselector 72, producing a position signal 88 having an amplitude R sin dP sin 0. This second position signal 88 is supplied to display device 35to control operation of the display conjointly with the first positionsignal 85.

In display apparatus 35A, detector 91 receives an output signal from theinternal resolver apparatus (not shown) of device 35 and utilizes thatsignal to control a track deviation pointer in the display device.Detector 92 receives another output signal from the same internalapparatus in device 35 and, in conjunction with servo apparatus 93,controls the distance-to-go indicator of the display device, showing thedistance to the selected way point as long as switch 94 is in theillustrated position. Detector 92 also actuates a to-from indicator (notshown) in display device 35 to inform the pilot whether he is flyingtoward or away from the selected waypoint. It will be recognized thatthe internal construction of the display apparatus has beensubstantially simplified because the mechanism and appropriate operatingcircuits are well known in the art.

The functional significance of the operations performed in computer 50in relation to the actual operation of an aircraft is illustrated inFIG. 3. In FIG. 3, the present position of an aircraft is indicated atpoint 101. The aircraft receives signals from a VORTAC (or TACAN)station located at point 102, the geographical orientation of the VORTACstation being indicated by phantom line 103 showing the direction ofmagnetic north and south.'The aircraft is to proceed from position 101to a waypoint 104, which may be an intermediate point on an extendedflight or may constitute the temiinus of the flight.

The direct path between aircraft position 101 and VOR- TAC station 102is indicated in FIG. 3 by line R. Infonnation as to the distance R isderived, in the aircraft, from the DME signals radiated by station 102and detected in the DME receiver 12 (FIG. 2). That is, the output of theDME receiver is a signal representative of the distance R but gives noindication of the angular orientation of the flight path that thisdistance represents. The bearing of the aircraft relative to station 102is indicated by the angle 4; in FIG. 3. This bearing information isderived, in the aircraft, by the VCR receiver circuits 1 1 and isrepresented by the phase displacement'between signals 41 and 44 (FIG.2).

From FIG. 3, it will be seen that the flight path R to the station 102can be represented by two linear coordinates X and Y, where X R sin dzY= R cos 4: From the foregoing description, however, it will be apparentthat these two linear coordinates correspond to the output signals ofthe initial computer portion of computer 50. That is, the Y coordinateis represented by the output signal from circuit 31 and the X coordinatefinds its counterpart in the output signal from circuit 32. Moreover,the same coordinate signals are represented by the outputs 83 and 86 ofintegrators 65 and 69 respectively, following modulation with the signalfrom oscillator 58.

The path fromstation 102 to waypoint 104, in FIG. 3, is represented bythe line P. Again, this path can be represented by two linearcoordinates X and Y,

Y P cos 0. In the computer of FIG. 2, it will be seen that these twolinear coordinates are present as the output signals 84 and 87 from thewaypoint selector 72. Moreover, like the output signals 83 and 86 frommodulator means 60 and 70, the waypoint coordinate signals 84 and 87 aremodulated with the signal from oscillator 58. V V

In FIG. 3, the path from the present aircraft position at point 101 tothe waypoint 104 is represented by line R". It will be immediatelyapparent that this path can also represented by two linear coordinates,

Y" =Rcos+Pcos 0. That is, the one linear coordinate X" is the sum of thelinear coordinates X and X and the other linear coordinate Y" is the sumof the coordinates Y and Y. But this is precisely the significance of vthe two output signals 85 and 88 from the summing amplifiers 66 and 71in the computer. Thus, the coordinate signals supplied to display device35 in the computer of FIG. 2 are directly representative of the linearcoordinates of the flight path R" required for the aircraft to travelcuit 51 is connected to the input of a constant slew rate or constantslope filter 102. Filter 102 is a feedback amplifier circuitincorporating a capacitor that is charged at a constant rate at alltimes, as long as the amplitude of the input signal is maintained abovea given minimum amplitude, producing an output signal 111 of triangularwaveform. Furthermore, this constant slew rate filter circuit afl'ords afixed phase shift, so that no phase errors are introduced in thecomputer circuit. A specific example of an appropriate filter circuit isdescribed hereinafter; see FIG. 6.

The output of filter 102 is applied to a sine wave synthesizer circuit103 to convert the triangular waveform signal 11 1 to a sinusoidalsignal 112. Synthesizer 103 is, essentially, a series ofdiode-controlled voltage dividers ina plural emitter-follower circuit,capable of producing an output signal of clean sine wave configurationin response to an input signal of triangular waveform. A specificexample of circuit 103 is described in connection with FIG. 7.

The output of synthesizer circuit 103 is supplied to a potentiometer104. The tap on potentiometer 104 is connected to the output shaft ofthe DME receiver 12 so that the amplitude of the output signal derivedfrom the potentiometer tap .is a function of the distance data developedby the DME receiver. The tap on potentiometer 104 is electricallyconnected to the input of an amplifier 105 and the output of amplifier105 is coupled to one input of the cosine multiplier detector 31 and toone input of the sine multiplier detector 32.

The output of zero crossing detector 57 (signal A) is connected to thesecond input of detector 31, as in the previously 7 describedembodiment. The output of detector 57 is also connected to the input ofa phase shift circuit 56A having its output connected to the secondinput of detector 32. Phase shifter circuit 56A is shown as comprisingan input circuit constituting a constant slew rate filter 106 and anoutput circuit comprising a zero crossing detector 107. It can bedemonstrated that the filter can be operated to produce a phase shift of90 as required for operation of sine detector 32. V The remainder ofFIG. 4 is essentially similar to the system of FIG. 2. It includes thetwo modulator means 60 and 70 connected to the outputs of detectors 31and 32 respectively. Modulator means 60 and 70 may be of the typedescribed above in-connection with FIG. 2, or may constitute any othersolid-state circuit capable of converting the DC. outputs of detectors31 and 32 to appropriate sinusoidal signals. The outputs of the twomodulator means 60 and 70, each of which has a secondary input fromlocal oscillator 58, are connected to the cosine summing amplifier 66and to the sine summing amplifier 71, respectively. Amplifier 71receives a second input signal p sin 6, from the waypoint selector 72.Amplifier 66 receives a second input signal, p cos 0, from the waypointselector 72. The outputs of the summing amplifiers are each connected todisplay apparatus 35A; the remaining connections for the displayapparatus may be as illustrated in FIG. 2.

Because the operation of computer is essentially similar to the systemsdescribed above, only a relatively brief description is necessary. Thevariable phase 30 hz. data signal appearing at output terminal 24 of theVCR receiver circuit unit 1 1 is applied to zero crossing detector 51 toproduce an output signal 81 of rectangular waveform. Signal 81 issupplied to the input of the constant slew rate filter 102, producing anoutput signal 111 of triangular waveform that is supplied to the diodesine wave synthesizer circuit 103. The output of circuit 103 is a sinewave signal 112 that is applied to the potentiometer 104. The positionof the tap on potentiometer 104 is varied in accordance with themechanical output of DME receiver 12, supplying a signal to amplifier105 that has a phase and frequency determined by the variable VOR datasignal and an amplitude determined by the DME data signal. This signalis applied to the cosine multiplier detector 31 where it is multipliedby the VCR reference signal that has been converted to rectangularwaveform in zero crossing detector '57 (signal 45A). The output frommultiplier 31 is a DC signal having an amplitude and polarityrepresentative of the vector distance R cos d: (see FIG. 3).

The square wave constant-phase VOR reference signal 45A from the outputof circuit 57 is also supplied to the constant slew rate filter 106,which produces an output signal 113 of triangular wavefonn that isshifted 90 in phase relative to the input signal. The triangularwaveform signal 113 is applied to the zero crossing detector 107 toproduce an output signal 46A of rectangular waveform. Signal 46A issupplied to one input of the sine multiplier detector circuit 32, theother input of detector 32 receiving the sinusoidal output signal fromamplifier 105. The output from detector 32 is a DC signal having anamplitude and polarity representative of the vector coordinate distanceR sin ti: (FIG. 3).

The DC output signals from detectors 31 and 32 are supplied to thecosine modulator means 60 and to the sine modu lator means 70,respectively. The locally generated carrier signal from oscillator 58 isalso supplied to the modulator means 60 and 70. The outputof modulatormeans 60 is a signal of sinusoidal waveform having an amplitude andphase representative of the vector distance R cos 5. The output ofmodulator means 70 is a sinusoidal signal having an amplitude and phaserepresentative of the coordinate vector distance R sin 4;.

waypoint selector 72, in FIG. 4, functions as in the circuit of FIG. 2to produce two output signals P cos and P sin 0, both at the localcarrier frequency of 465 hz. The waypoint vector signal P sin 0 issupplied to the summing amplifier 71 together with the output signalfrom modulator means 70, producing a waypoint coordinate signal R sin dzP sin 0 that is supplied to the display apparatus 35A. Similarly, thewaypoint vector signal P cos 0 is supplied to the summing amplifier 66along with the output signal from modulator means 60, producing awaypoint coordinate signal R cos d P cos 0 that is supplied to thedisplay apparatus 35A.

FIG. illustrates, in schematic form, a typical zero crossing detectorthat may be used as circuit 51 in computer 100 (FIG. 4). The circuitillustrated in FIG. 5 is also appropriate for use as the circuit 57 orthe circuit 107 of FIG. 4.

Referring to FIG. 5, the zero crossing detector illustrated thereincomprises an operational amplifier 151 having its inverting inputterminal connected through a resistor 152 to a source of a sinusoidalsignal, shown in FIG. 5 as the output terminal 24 of the VCR receiverunit. The inverting input terminal of operational amplifier 151 is alsoreturned to ground through a diode 153 and is connected to one terminalof a diode bridge 154. The opposite terminal of the diode bridge isconnected to the output terminal of operational amplifier 151. Oneintermediate terminal of the diode bridge is connected through aresistor 155 to a negative power supply designated as V. The remainingterminal of the diode bridge is connected through a resistor 156 and adiode 157 to a positive power supply designated as V+.

The non-inverting input terminal of the operational amplifier 151 isconnected to ground through a resistor 158. The two power supplyterminals of the operational amplifier are connected to the V+ and V-supplies. The output terminal of the operational amplifier is connectedthrough the parallel combination of a resistor 161 and a capacitor 162to the base electrode of a transistor 163.

The emitter of transistor 163 is connected to system ground. Thecollector of transistor 163 is connected through the series combinationof a resistor 164 and a resistor 165 to the V+ supply. The commonterminal of resistors 164 and is connected to the base of a secondtransistor 166. The emitter of transistor 166 is connected to the V+supply and the collector is connected through a resistor 167 to systemground. The output terminal 168 for the circuit is connected to thecollector of transistor 166.

When the sinusoidal signal 41 is applied to the inputterminal 24 of thezero crossing detector, as illustrated in FIG. 5,

the output signal appearing at terminal 168 is the square wave signal 81having a peak positive amplitude of V+ and a peak negative amplitude ofV-, the square wave signal being centered around zero. Substantialvariations in amplitude of the input signal 41 can be tolerated by thecircuit with no variation in the amplitude of the output signal 81.Furthermore, zero crossing detector 51, as shown in FIG. 5 does notintroduce any perceptible phase shift in the output signal as comparedwith the input signal. This characteristic is maintained despitevariations in the power supply and other environmental factors. Thus,one of the major sources of difficulty in aircraft radio navigationsystems, the introduction of errors in the receiver and signalprocessing apparatus, is effectively eliminated insofar as this circuitis concerned.

FIG. 6 illustrates a constant slew rate filter circuit that may beutilized as the filter'102 in the system of FIG. 4. By the same token,the circuit shown in FIG. 6 can be employed as v the filter 106 in thesystem of FIG. 4.

The constant slew rate filter 102, as illustrated in FIG. 6, comprisesan operational amplifier 171 having its non-inverting input terminalconnected through a resistor 172 to the output terminal 168 of thepreceding stage of the computer (see FIG. 5). The power supply terminalsof operational amplifier 171 are connected to the V+ and V-- supplies.The output terminal 173 of the operational amplifier, which is also theoutput terminal for the complete filter circuit, is connected to acapacitor 174 that is in turn connected to system ground through a smallresistor 175. The common terminal of capacitor 174 and resistor 175 isconnected back to the inverting input of operational amplifier 171.

The output terminal 173 of operational amplifier 171 is also connectedback to the non-inverting input of the operational amplifier 171 by a DCand low-frequency stabilizing feedback circuit 176. Feedback circuit 176includes a pair of oppositely-oriented diodes 177 and 178 connected tooutput terminal 173. Diode 177 is returned to system ground through aparallel RC circuit comprising a resistor 181 and a capacitor 182.Similarly, diode 178 is returned to ground through a parallelcombination of a resistor 183 and a capacitor 184. In addition, diodes177 and 178 are connected to the end terminals of a potentiometer 185.

The tap on potentiometer 185 is connected to a resistor 186 that isconnected to the inverting input terminal of an operational amplifier187. The power supply terminals of the operating amplifier 187 areconnected to the V+ and V- supplies, respectively. The non-invertinginput terminal to operational amplifier 187 is connected to groundthrough a resistor 188. The output terminal of operational amplifier 187is connected through a feedback resistor 189 to the inverting input ofthe amplifier.

The output of operational amplifier 187 is connected to a resistor 191that is in turn connected to the center terminal of a voltage dividercomprising a resistor 192 that is connected to system ground and aresistor 193 that is connected back to the non-inverting input ofoperational amplifier 171.

The basic operation of the constant slew rate filter circuit illustratedin FIG. 6 can be most easily considered by initially avoiding referenceto the stabilization feedback circuit 176. A reference voltage (ofeither polarity) applied to the input terminal 168 of the filter circuitis amplified and appears, without change of polarity, at the outputterminal 173. This voltage charges capacitor 174 through the lowresistance afforded by resistor 175. The voltage drop across resistor175, which is a function of the charging current through capacitor 174,is fed back to the inverting input terminal of the operational amplifier171. The output voltage of amplifier 171 will be such that the voltageacross resistor 175 precisely matches the input voltage.

With constant voltage drop across resistance 175, the charging rate ofcapacitor 174 is a straight line function. That is, the charging rate ofcapacitor 174 has a constant slope, so that the circuit may beconsidered to have a constant slewing rate". The actual slewing rate isdetermined by the impedance of capacitor 174 and resistor 175 and theamplitude of the input voltage. The direction of the slope of thecharging characteristic for capacitor 174 depends, of course, on thepolarity of the input signal. Furthermore, it can be demonstrated thatthe phase shift of the circuit is precisely 90 at any frequency above agiven cut-off or corner frequency determined by the impedances of thecircuit, particularly capacitor 174 and resistor 175.

The basic constant slope or constant slew rate filter, as thus faroperationally described, is not stabilized for DC excitation.Consequently, there is a tendency for the output voltage at terminal 173to drift up to the value of 1+ or down to V, depending upon variationsin the DC content in the input signal and other factors. Thischaracteristic can also produce fluctuations, at extremely lowfrequencies, in the output signal 111. The DC stabilization feedbackcircuit 176, which can be varied considerably from the form shown,prevents this DC drift.

FIG. 7 illustrates one form of sine wave synthesizer that may beutilized as the circuit 103 in the system of FIG. 4. In FIG. 7, theinput terminal 173, which is the output terminal for the precedingconstant slew rate filter, is connected to a resistor 201 that is inturn connected to an output terminal 261. The remaining components ofthe synthesizer are all connected to output terminal 261.

Thus, output terminal 261 isconnected through a diode 221 to a series ofprecision resistors 202, 203, 204, 205 and 206, with resistor 206 beingconnected through a resistor 207 to the V- supply. Output terminal 261is also connected through the series combination of a resistor 208 and adiode 222 to the common terminal of resistors 202 and 203, through aresistor 209 and a diode 223 to the common terminal of resistors 203 and204, through a resistor 210 and a diode 224 to the common terminal ofresistors 204 and 205, through a resistor 211 and a diode 225 to thecommon terminal of resistors 205 and 206, and through a resistor 212 anda diode 226 to the common terminal of resistors 206 and 207. The lowerhalf of the synthesizer comprises a diode 227 that connects the outputterminal 261 to a series of precision resistors 213, 214, 215, 216 and217, resistor 217 being connected through a resistor 218 to the V+supply. The junctions of the resistors in this series 213-217 areconnected to the resistors 208, 209, 210, 211 and 212 by individualdiodes 228, 229, 230 and 231 and 232 respectively.

Diode 221, in addition to its connection to resistor 202, is connectedto the emitter of a transistor 241 and to a blocking capacitor 219 thatis returned to the V- supply. The collector of transistor 241 isconnected to the V+ power supply. The base of the transistor isconnected to system ground through a resistor 245 and is also connectedto the V+ supply through the series combination of a resistor 246 and avariable resistor 247. Similarly, diode 227 is connected to the emitterof a transistor 242. The collector of transistor 242 is connected to theV supply and the base of the transistor is connected to ground through aresistor 248 and to the V- supply through the series combination of aresistor 249 and a variable resistor 251.

At the right-hand side of the synthesizer circuit of FIG. 7, diode 226is connected to the emitter of a transistor 243 having its collectorconnected to the V+ supply. The base of transistor 243 is connected toground through a resistor 252 and is connected to the V+ supply througha resistor 253 and a variable resistor 254. Similarly, diode 232 isconnected to the emitter of a transistor 244 having its collectorconnected to the V- supply. The base of transistor 244 is connected toground through a resistor 255 and is also connected to the V- supplythrough the series combination of a resistor 256 and a variable resistor257.

Output terminal 261 is.also connected to the V+ supply through aresistor 262. In addition, the output terminal is connected to aresistor 263 in turn connected to the non-inverting input of anoperational amplifier 264. The power supply terminals of operationalamplifier 264 are connected to the V+ and V- supplies. The outputterminal of operational amplifier 264 is connected to the DME-controlledpotentiometer 104, which is returned to system ground. In addition, theoutput terminal of the operational amplifier is connected to a feedbackresistor 265 that is returned to the inverting input of the amplifier.

Synthesizer 103, as illustrated in FIG. 7, is a generally known circuitthat has been utilized in test equipment; accordingly, no detailedoperational description for the synthesizer is provided herein. Usingselected resistors of appropriate ly graduated size, and with anadequate number of stages as illustrated in FIG. 7, the synthesizercircuit produces a sine wave output signal 112, having virtually noharmonic distor tion, from the triangular waveform input signal 11 1.

FIG. 8 illustrates specific circuits that may be utilized for amplifier105, cosine multiplier detector 31, and sine multiplier detector 32 inthe computer system of FIG. 4. At the lefthand side of FIG. 8, theDME-actuated potentiometer 104 is shown, the tap on the potentiometerbeing connected through a resistor 301 to the non-inverting input of anoperational amplifier 312. The power supply terminals of the operationalamplifier are connected to the V+ and V supplies in the usual manner.The output terminal 313 of the operational amplifier is connected to afeedback resistor 314 that is in turn connected to the inverting inputof the amplifier.

The circuit for the cosine multiplier detector 31 that is illustrated inFIG. 8 is, essentially, a field-effect transistor chopper. It comprisesa field effect transistor 316 having one main electrode connected tosystem ground and having the other main electrode connected to aresistor 317 that is connected to the output terminal 313 of amplifier105. The gate electrode of transistor 316 is connected to a resistor 318in a circuit that extends back to the zero crossing detector 57 in thereference signal channel of the receiver (see FIG. 4). The outputcircuit of the cosine multiplier detector 31, as shown in FIG. 8,comprises a conventional low-pass filter including two series resistors319 and 321 and two shunt capacitors 322 and 323.

The circuit illustrated in FIG. 8 for the sine multiplier detector 32 isessentially the same as for the cosine detector 31. It includes afield-effect transistor 325 having one main electrode connected tosystem ground; the other main electrode is connected to system ground;the other main electrode is connected to a resistor 326 that isconnected back to the output terminal 313 of amplifier 105. The secondinput to the detector comprises a resistor 327 connected from zerocrossing detector 107 (FIG. 4) to the gate electrode of transistor 325.The output of detector 32 is a low pass filter comprising two seriesresistors 328 and 329 and two shunt capacitors 331 and 332.

The operation of amplifier is quite conventional and need not bedescribed in detail. It supplies a sinusoidal signal to the field-effecttransistor chopper circuit of detector 31, transistor 316 being gated onand off in alternate half cycles of the square wave input signal 45Afrom zero crossing detector 57. The output signal is a pulsating DCsignal having a polarity that varies in accordance with the phaserelationship between signals 45A and 112 and having an amplitudedetermined by the amplitude of signal 112. The waveform of the signalappearing at chopper terminal 334 is indicated by the solid linewaveform 335 in FIG. 8A, for conditions in which the phase angle betweenthe two input signals 45A and 112 is zero degrees. Theoutput signalappearing at terminal 336 is represented by the dash. line signal 337,in FIG. 8A, for the same operating conditions.

The operation of the sine multiplier detector 32 of FIG. 8A is the sameas described above for detector 31. The solid line waveform 340 in FIG.8A represents the signal developed at the chopper terminal 338 forcircuit conditions corresponding to a phase difference of 90 between theinputsignals 112 and 337 and 341 change in amplitude andin polarity aswith changes in the relativephase angles of the two input signals toeach detector; signals 337 and 341 also vary in amplitude with changesin the amplitude of the sinusoidal input signal 112.

FIG. 9 illustrates one apparatus that can be utilized for the waypointselector 72. It includes an input potentiometer 401 having one terminalconnected to .the oscillator 58 (P10. 4) and the other terminal returnedto system ground. The position of the tap on potentiometer 401 isadjusted to vary the amplitude of the output signal from the tap tocorrespond to the distance P from the ground station to the" selectedwaypoint (see FIG. 3).

The tap on potentiometer 401 is connected to the input of an amplifier402 and the output of an amplifier 402 is connected to the input winding404 of a rotary resolver 403. Resolver 403 includes two orthagonaloutput windings 405 and 406 connected in a conventional arrangement andhaving output terminal 407 and 408 respectively. The angular orientationbetween resolver input winding 404 and the output windings 405 and 406is manually adjusted to correspond to the bearing angle 9 for thewaypoint (see FIG. 3).,

In operation, the 465 hz. signal from oscillator 58 is adjusted inamplitude by potentiometer 401 and in phaseby resolver 403. In this way,two output signals are produced, at terminals 407 and 408, of the form Psin and P cos 0, respectively. It should be understood that thisconstruction for waypoint selector 72 is presented only in the interestof completeness of disclosure and that other appropriate waypoint signalgenerating devices, such as multi-tap adjustable transformers, can beemployed with the computer of the present invention in the generation ofwaypoint coordinate signals.

FIG. 10 illustrates an appropriate summing amplifier that may be usedfor the amplifier circuit 66 or for the amplifier circuit 71 in FIG. 4.As shown in FIG. 10, the sinusoidal ground station coordinate signal Rcos 4: is applied to one end of a circuit comprising a resistor 411, apotentiometer 412, and a resistor 413 all connected in series with eachother. The

other end of this same circuit is supplied with the sinusoidal waypointcoordinate signal P cos 0. The tap on potentiometer 412 is connected tothe input of an amplifier 414 and the output signal is of the form R cos4: cos 6.

ln order to afford a more complete illustration of the invention,certain circuit parameters are set forth in detailed tabularformhereinafter. It should be understood that this information ispresented solely by way of example and in no sense as a limitation ofthe invention.

317,326 100 kilohms 20s 7 432 ohms 209' 1370 ohms 210 1780 ohms 2ll 4530ohms 212 6650 ohms 3 l9,328 500 ohms 321,329 5 kilohms CAPACITORS 1620.1 microfarads I74 1 microfarad l82,l84 4 microfarad 2 l9 3.3microfarads Semi-Conductor Devices 163,244,243 2N2222 l66,242,244 2N2907[77,178 lN270 22l through 232 lN9l6B 3l6,325 2N4222 1 Voltage SuppliesV+ +|2 volts v l2 volts lclaim:

" 1. ln an aircraft navigation system including receiver means fordeveloping first and second data signals of given frequency varying inphase angle relative to each other, in accordance with one positionparameter, and a third data signal representative of another positionparameter, relative to a ground station transmitter, an all-electronicsolid state analog position computer comprising:

means utilizing said first data signal to generate a first intermediatesignal, at said given frequency, of substantially rectangular wave form;

modulating means for amplitude modulating said second data signal inaccordance with said third data signal to generate a second intermediatesignal of amplitude R, at said given frequency, having a substantiallysinusoidal wave form;

a first product detector for multiplying said intermediate signalstogether to generate a first ground station coordinate signal R cos da;

and a second product detector for multiplying said intermediate signalstogether, with one intermediate signal shifted in phase, to generate asecond ground station coordinate signal R sin 2. An aircraft navigationposition computer according to claim 1 in which said modulating meanscomprises a square wave signal generator, driven by said second datasignal to control the frequency and phase of its output and driven bysaid third data signal to control the amplitude of its output, and twosuccessive integrator stages coupled to the output of said square wavesignal generator to convert its square wave output to said secondintermediate signal.

3. An aircraft navigation position computer according to claim 2 inwhich said square wave signal generator includes a zero crossingdetector actuated by said second data signal and a gated chopper circuithaving its chopping rate controlled by the output of said zero crossingdetector and further having its amplitude controlled by said third datasignal.

4. An aircraft navigation position computer according to claim 1 inwhich said modulating means for generating said secondintennediatesignal comprises, in series,

a zero crossing detector, driven by said second data signal, forgenerating a square wave signal corresponding in frequency and phase tosaid second data signal;

a constant slew rate filter for converting said square wave signal to asignal of triangular wave form;

a sine wave synthesizer for converting said signal of triangular waveform to a sinusoidal signal of constant amplitude;

and a variable output circuit for said sine wave synthesizer,continuously varied in accordance with said third data signal.

5. An aircraft navigation position computer according to claim 1 inwhich said means for generating said first intermediate signal is a zerocrossing detector actuated by said first data signal,

said computer further including phase shifting means comprising aconstant slew rate filter driven by said first intermediate signal and azero crossing detector actuated by the output of said filter to afford aphase shift of 90 in said first intermediate signal for utilization insaid second product detector.

6. An aircraft navigation position computer according to claim 1 andfurther comprising:

a waypoint selector for generating first and second waypoint coordinatesignals P cos 6 and P sin representative of the location of a selectedwaypoint relative to the ground station;

a first summing amplifier for algebraically combining said first groundstation coordinate signal and said first waypoint coordinate signal todevelop a first linear coordinate position signal R cos (b P cos 0; andsecond summing amplifier for algebraically combining said second groundstation coordinate signal and said second waypoint coordinate signal todevelop a'second linear coordinate position signal R sin (b P sin 6.

7. An aircraft navigation position computer according to claim 6 inwhich said waypoint selector is actuated by a carrier signal from alocal oscillator and develops waypoint coordinate signals of sinusoidalwaveform at a given low carrier frequency;

said computer comprising sine modulating means and cosine modulatingmeans for modulating said ground station coordinate signals with saidcarrier signal to produce ground station coordinate signals ofsinusoidal waveform prior to combination of said ground stationcoordinate signals with said waypoint coordinate signals in said summingamplifiers.

8. An aircraft navigation position computer according to claim 7 inwhich said carrier signal, as applied to said sine and cosine modulatingmeans, is of rectangular wave form, in which said further modulatingmeans includes two signal channels, each comprising an amplitudemodulator for modulating said square wave carrier signal with one ofsaid ground station coordinate signals, followed by two successiveintegrators for developing said ground station coordinate signals ofsinusoidal waveform.

9. The method of generating a continuous, precise trigonometric functionsignal based upon data signals of given frequency having a phasedisplacement (1:, relative to each other, that is representative of thefunction angle, comprising:

processing one data signal to develop a first intermediate signal, atsaid given frequency, of relatively pure sinusoidal waveform; squaringthe other data signal to develop a second intermediate signal, at saidgiven frequency, having a fast risetime waveform of essentiallyrectangular configuration;

chopping the first intermediate signal with the second intermediatesignal to produce a pulsating DC signal having an amplitude proportionalto the cosine of the phase angle between said data signals; V 3

and filtering said pulsating DC signal to develop a steady DC outputsignal having an amplitude continuously representative of said phaseangle dz.

10. The method of generating a pair of continuous, precise sine andcosine function signals based upon first and second data signals ofgiven frequency having a phase displacement 4:, relative to each other,that is representative of a function angle, comprising:

processing the first data signal to develop a first intermediate signal,at said given. frequency, of relatively pure sinusoidal waveform;squaring the second data signal to develop a second intermediate signal,at said given frequency, having a fast risetime waveform of essentiallyrectangular configuration;

shifting the phase of the second intermediate signal by and re-squaringto develop a third intermediate signal. at said given frequency, havinga fast rise-time waveform of essentially rectangular configuration andin phase quadrature with said second intermediate signal;

chopping the first intermediate signal with the second intermediatesignal and the third intermediate signal, in separate chopping circuits,to produce two pulsating DC signals, one having an amplitudeproportional to the cosine of the phase angle between the first andsecond data signals and the other being an amplitude proportional to thesine of the phase angle between data signals;

and filtering said pulsating DC signals to develop steady DC outputsignals having amplitudes continuously representative of the cosine andsine, respectively, of said phase angle 4:.

11. The method aircraft navigation, entailing conversion of navigationdata from polar to rectilinear coordinates, using first and second ACdata signals of given frequency having a phase displacement 4), relativeto each other, that is representative of the polar coordinate angle, anda third data signal comprising a DC signal having an amplitude Rproportional to the polar coordinate vector length, comprising:

squaring the first data signal;

chopping the third data signal with the squared firstdata signal toproduce a variable amplitude signal of amplitude v R, at said givenfrequency, of essentially rectangular waveform; filtering the variableamplitude signal to develop a first intermediate signal, at said givenfrequency, of relatively pure sinusoidal waveform; squaring the seconddata signal to develop a second intermediate signal, at said givenfrequency, having a fast risetime waveform of essentially rectangularconfiguration; chopping the first intermediate signal with the secondintermediate signal to produce a pulsating DC signal having an amplitudeproportional to R cos (b: and filtering said pulsating DC signal todevelop a steady DC coordinate signal Y= R cos d). 12. The method ofaircraft navigation, according to claim 1 1, and further comprising:

phase-shifting the second data signal, and then re-squaring, to developa third intermediate signal, at said given frequency, of essentiallyrectangular waveform, in phase guadrature to said second intermediatesignal; chopping the first intermediate signal with the thirdintermediate signal to produce a second pulsating DC signal having anamplitude proportional to R sin o; and filtering said second pulsatingDC signal to develop a second steady DC coordinate signal X ==R sin d:13. The method-of aircraft navigation according to claim 12, and furthercomprising:

generating waypoint coordinate signals X =P sin Qind Y P cos 0representative of the position of a waypoint relative to the source ofsaid data signals; and summing said waypoint coordinate signals withsaid steady DC coordinate signals to develop rectilinear coordinateposition signals X"=Rsin 4 +Psin0and Y"=Rcos+Pcos0. 14. The method ofaircraft navigation according to claim 13, and further comprising:

instrument display.

1. In an aircraft navigation system including receiver means fordeveloping first and second data signals of given frequency varying inphase angle phi relative to each other, in accordance with one positionparameter, and a third data signal representative of another positionparameter, relative to a ground station transmitter, an all-electronicsolid state analog position computer comprising: means utilizing saidfirst data signal to generate a first intermediate signal, at said givenfrequency, of substantially rectangular wave form; modulating means foramplitude modulating said second data signal in accordance with saidthird data signal to generate a second intermediate signal of amplitudeR, at said given frequency, having a substantially sinusoidal wave form;a first product detector for multiplying said intermediate signalstogether to generate a first ground station coordinate signal R cos phi; and a second product detector for multiplying said intermediatesignals together, with one intermediate signal shifted 90* in phase, togenerate a second ground station coordinate signal R sin phi .
 2. Anaircraft navigation position computer according to claim 1 in which saidmodulating means comprises a square wave signal generator, driven bysaid second data signal to control the frequency and phase of its outputand driven by said third data signal to control the amplitude of itsoutput, and two successive integrator stages coupled to the output ofsaid square wave signal generator to convert its square wave output tosaid second intermediate signal.
 3. An aircraft navigation positioncomputer according to claim 2 in which said square wave signal generatorincludes a zero crossing detector actuated by said second data signaland a gated chopper circuit having its chopping rate controlled by theoutput of said zero crossing detector and further having its amplitudecontrolled by said third data signal.
 4. An aircraft navigation positioncomputer according to claim 1 in which said modulating means forgenerating said second intermediate signal comprises, in series, a zerocrossing detector, driven by said second data signal, for generating asquare wave signal corresponding in frequency and phase to said seconddata signal; a constant slew rate filter for converting said square wavesignal to a signal of triangular wave form; a sine wave synthesizer forconverting said signal of triangular wave form to a sinusoidal signal ofconstant amplitude; and a variable output circuit for said sine wavesynthesizer, continuously varied in accordance with said third datasignal.
 5. An aircraft navigation position computer according to claim 1in which said means for generating said first intermediate signal is azero crossing detector actuated by said first data signal, said computerfurther including phase shifting means comprising a constant slew ratefilter drivEn by said first intermediate signal and a zero crossingdetector actuated by the output of said filter to afford a phase shiftof 90* in said first intermediate signal for utilization in said secondproduct detector.
 6. An aircraft navigation position computer accordingto claim 1 and further comprising: a waypoint selector for generatingfirst and second waypoint coordinate signals P cos theta and P sin thetarepresentative of the location of a selected waypoint relative to theground station; a first summing amplifier for algebraically combiningsaid first ground station coordinate signal and said first waypointcoordinate signal to develop a first linear coordinate position signal Rcos phi + P cos theta ; and a second summing amplifier for algebraicallycombining said second ground station coordinate signal and said secondwaypoint coordinate signal to develop a second linear coordinateposition signal R sin phi + P sin theta .
 7. An aircraft navigationposition computer according to claim 6 in which said waypoint selectoris actuated by a carrier signal from a local oscillator and developswaypoint coordinate signals of sinusoidal waveform at a given lowcarrier frequency; said computer comprising sine modulating means andcosine modulating means for modulating said ground station coordinatesignals with said carrier signal to produce ground station coordinatesignals of sinusoidal waveform prior to combination of said groundstation coordinate signals with said waypoint coordinate signals in saidsumming amplifiers.
 8. An aircraft navigation position computeraccording to claim 7 in which said carrier signal, as applied to saidsine and cosine modulating means, is of rectangular wave form, in whichsaid further modulating means includes two signal channels, eachcomprising an amplitude modulator for modulating said square wavecarrier signal with one of said ground station coordinate signals,followed by two successive integrators for developing said groundstation coordinate signals of sinusoidal waveform.
 9. The method ofgenerating a continuous, precise trigonometric function signal basedupon data signals of given frequency having a phase displacement phi ,relative to each other, that is representative of the function angle,comprising: processing one data signal to develop a first intermediatesignal, at said given frequency, of relatively pure sinusoidal waveform;squaring the other data signal to develop a second intermediate signal,at said given frequency, having a fast rise-time waveform of essentiallyrectangular configuration; chopping the first intermediate signal withthe second intermediate signal to produce a pulsating DC signal havingan amplitude proportional to the cosine of the phase angle between saiddata signals; and filtering said pulsating DC signal to develop a steadyDC output signal having an amplitude continuously representative of saidphase angle phi .
 10. The method of generating a pair of continuous,precise sine and cosine function signals based upon first and seconddata signals of given frequency having a phase displacement phi ,relative to each other, that is representative of a function angle,comprising: processing the first data signal to develop a firstintermediate signal, at said given frequency, of relatively puresinusoidal waveform; squaring the second data signal to develop a secondintermediate signal, at said given frequency, having a fast rise-timewaveform of essentially rectangular configuration; shifting the phase ofthe second intermediate signal by 90* and re-squaring to develop a thirdintermediate signal, at said given frequency, having a fast rise-timewaveform of essentially rectangular configuration and in phasequadrature with said second intermediate signal; chopping the firstintermediate signal with the second intermediate signal and the thirdintermediate signal, in separate chopping circuits, to produce twopulsating DC signals, one having an amplitude proportional to the cosineof the phase angle between the first and second data signals and theother being an amplitude proportional to the sine of the phase anglebetween data signals; and filtering said pulsating DC signals to developsteady DC output signals having amplitudes continuously representativeof the cosine and sine, respectively, of said phase angle phi .
 11. Themethod aircraft navigation, entailing conversion of navigation data frompolar to rectilinear coordinates, using first and second AC data signalsof given frequency having a phase displacement phi , relative to eachother, that is representative of the polar coordinate angle, and a thirddata signal comprising a DC signal having an amplitude R proportional tothe polar coordinate vector length, comprising: squaring the first datasignal; chopping the third data signal with the squared first datasignal to produce a variable amplitude signal of amplitude R, at saidgiven frequency, of essentially rectangular waveform; filtering thevariable amplitude signal to develop a first intermediate signal, atsaid given frequency, of relatively pure sinusoidal waveform; squaringthe second data signal to develop a second intermediate signal, at saidgiven frequency, having a fast rise-time waveform of essentiallyrectangular configuration; chopping the first intermediate signal withthe second intermediate signal to produce a pulsating DC signal havingan amplitude proportional to R cos phi : and filtering said pulsating DCsignal to develop a steady DC coordinate signal Y R cos phi .
 12. Themethod of aircraft navigation, according to claim 11, and furthercomprising: phase-shifting the second data signal, and then re-squaring,to develop a third intermediate signal, at said given frequency, ofessentially rectangular waveform, in phase guadrature to said secondintermediate signal; chopping the first intermediate signal with thethird intermediate signal to produce a second pulsating DC signal havingan amplitude proportional to R sin phi ; and filtering said secondpulsating DC signal to develop a second steady DC coordinate signal X Rsin phi .
 13. The method of aircraft navigation according to claim 12,and further comprising: generating waypoint coordinate signals X'' P sintheta and Y'' P cos theta representative of the position of a waypointrelative to the source of said data signals; and summing said waypointcoordinate signals with said steady DC coordinate signals to developrectilinear coordinate position signals X'''' R sin phi + P sin thetaand Y'''' R cos phi + P cos theta .
 14. The method of aircraftnavigation according to claim 13, and further comprising: rotating saidrectilinear coordinate position signals to resolve cross track and alongtrack signal information for instrument display.